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MIC5190 micrel MIC5190 ultra high-speed, high-current active filter/ldo controller general description the MIC5190 is an ultra high-speed linear regulator. it uses an external n-channel fet as its power device. the MIC5190 offers ultra high-speed to cope with the fast load demands of microprocessor cores, asics, and other high-speed devices. signal bandwidths of greater than 500khz can be achieved with a minimum amount of capacitance while at the same time keeping the output voltage clean, regardless of load demand. a powerful output driver delivers large mosfets into their linear regions, achieving ultra-low dropout voltage. 1.25v in 10% can be turned into 0.9v 1% without the use of a large amount of capacitance. MIC5190 (0.5v reference) is optimized for output voltages of below 1.0v. the MIC5190 is offered in 10-lead 3mm 3mm mlf? and 10-lead msop-10 packages and has an operating junction temperature range of C40 c to +125 c. all support documentation can be found on micrels web site at www.micrel.com. typical application features ? input voltage range: v in = 0.9v to 5.5v ? +1.0% initial output tolerance ? dropout down to 25mv@10a ? filters out switching frequency noise on input ? very high large signal bandwidth >500khz ? psrr >40db at 500khz ? adjustable output voltage down to 0.5v ? stable with any output capacitor ? excellent line and load regulation specifications ? logic controlled shutdown ? current limit protection ?3mm 3mm 10-lead mlf? and msop-10 packages ? available C40 c to +125 c junction temperature applications ? distributed power supplies ? asic power supplies ? dsp, p, and c power supplies micrel, inc. ?2180 fortune drive ?san jose, ca 95131 ?usa ?tel + 1 (408) 944-0800 ?fax + 1 (408) 474-1000 ?http://www.mic rel.com microleadframe and mlf are trademarks of amkor technology, inc. powerpak is a trademark of siliconix, inc. vin sgnd pgnd is ir3716s out vcc1 r1 100? c3 0 . 01 f c 2 10 f r 2 1 25 ? fb gnd v out = 0 .9 v @ 7 a gnd vcc 2 e n co m p MIC5190 r3 1 2.5k ? c1 0 . 01 f v in = 1 .2 v v cc = 1 2 v december 2005 1 m9999- 120105
m i c 5 1 9 0 micrel pin description pin numbe r pin nam e pin function 1 vi n i nput voltage ( current sense +). 2 f b feedback input to error amplifier. 3 sgn d s ignal ground. 4 vcc 1 s upply to the internal voltage regulator. 5 co m p error amplifier output for external compensation . 6 e n enable ( i nput) : c m o s -compatible . logic high = enable, logic low = s hutdown . d o not float pin . 7 vc c 2 p ower to output driver . 8 o u t o utput drive to gate of power m o s f e t . 9 pgn d p ower ground . 1 0 i s c urrent sense . pin configuration co m p e n 5 1 vin fb sgnd vcc1 1 0 is pgnd out vcc 2 9 8 7 2 3 4 6 mlf ? -10 (ml) ordering information co m p e n 6 5 1 vin fb sgnd vcc1 1 0 is pgnd out vcc 2 9 8 7 2 3 4 msop-10 (mm) fb output output standard pb-free v oltage current v oltage MIC5190bml MIC5190yml 0.5v adj adj C40c to +125c 10-pin mlf? MIC5190bmm MIC5190ymm 0.5v adj adj C40c to +125c msop-10 part number package junction temp. range december 2005 2 m9999- 120105
m ic 5 1 9 0 micrel electrical characteristics (6) t a = 25 c with v in = 1 .2 v , v cc = 1 2 v , v out = 0 .5 v ; bold values indicate C4 0 c < t j < + 1 25 c ; unless otherwise specified. parameter condition min typ max units o utput v oltage accuracy at 25 c C 1 + 1 % o ver temperature range 2+2 % o utput v oltage line r egulation v in = 1 .2 v to 5.5 v C 0 . 10 . 00 5+ 0 . 1 %/ v feedback v oltage 0 .495 0 .5 0 .5 0 5 v o utput v oltage load r egulation i l = 10 ma to 1 a 0 . 0 2 0 .5 % vcc p in c urrent ( v cc 1 + v cc 2) enable = 0v 4 0 a vcc p in c urrent ( v cc 1 + v cc 2) enable = 5 v1 5 20 ma vin p in c urrent c urrent from v in 10 15 a fb bias c urrent 13 30 a c urrent limit t hreshold 35 5 0 70 m v s tart-up t ime v e n = v in 25 100 s enable i nput t hreshold r egulator enable 0.8 0 . 6v r egulator shutdown 0 .5 0.2 v enable hysteresis 100 m v enable p in i nput c urrent v i l < 0 .2 v ( r egulator shutdown) 100 na v i h > 0 .8 v ( r egulator enabled) 100 na notes: 1 . exceeding the absolute maximum ratings may damage the device. 2. d evices are e sd sensitive. handling precautions recommended. human body model, 1 .5k in series with 100 pf. 3 . t he device is not guaranteed to function outside its operating ratings. 4. p er je sd 5 1 -5 ( 1s 2 p d irect attach method). 5. p er je sd 5 1 - 3 ( 1s0p ). 6 . s pecification for packaged product only. absolute maximum ratings (1) s upply v oltage ( v in ) .................................................. + 6 . 0v enable i nput v oltage ( v e n ) ......................................... + 1 4 v v cc 1 , v cc 2 ............................................................... + 1 4 v junction t emperature ( t j ) ................ C4 0 c t j + 1 25 c e sd ......................................................................... note 2 operating ratings (3) s upply v oltage ( v in ) ................................... + 0 .9 v to +5.5 v enable i nput v oltage ( v e n ) ................................. 0v to v cc v cc 1 , v cc 2 ............................................... +4.5 v to + 13 .2 v junction t emperature ( t j ) ................ C4 0 c t j + 1 25 c p ackage t hermal r esistance mlf? ( ja ) (4) ..................................................... 60 c /w m sop ( ja ) (5) .............................................................. 2 00 c /w december 2005 3 m9999-120105
m ic 5 1 9 0 micrel typical characteristics 0 .495 0 .49 6 0 .49 7 0 .498 0 .499 0 .5 0 .5 01 0 .5 0 2 0 .5 03 0 .5 0 4 0 .5 0 5 01 2 3 45 67 89 10 o utput v oltage ( v ) o utput c urrent (a) load re g ulation 0 .2 0 . 3 0 .4 0 .5 0 . 6 0 . 7 0 .8 4.5 5.5 6 .5 7 .5 8.5 9.5 10 .5 11 .5 1 2.5 13 .5 e n t h ( v ) v cc v oltage ( v ) enable threshold vs. v cc voltage 0 2 4 6 8 10 1 2 1 4 16 1 8 2 0 -4 0 -2 00 2 0 4 060 8 0 100 1 2 0 i nput c urrent ( a) t emperature ( c ) input current vs. temperature 9 10 11 1 2 13 1 4 1 5 4.5 5.5 6 .5 7 .5 8.5 9.5 10 .5 11 .5 1 2.5 13 .5 feedback c urrent ( a) v cc v oltage( v ) feedback current vs. v cc voltage 0 5 10 1 5 2 0 25 -4 0 -2 00 2 0 4 060 8 0 100 1 2 0 feedback c urrent ( a) t emperature ( c ) feedback current vs. temperature 0 .495 0 .49 6 0 .49 7 0 .498 0 .499 0 .5 0 .5 01 0 .5 0 2 0 .5 03 0 .5 0 4 0 .5 0 5 4.5 5.5 6 .5 7 .5 8.5 9.5 10 .5 11 .5 1 2.5 v out ( v ) v cc ( v ) v out vs. v cc voltage 0 .495 0 .49 6 0 .49 7 0 .498 0 .499 0 .5 0 .5 01 0 .5 0 2 0 .5 03 0 .5 0 4 0 .5 0 5 -4 0 -2 00 2 0 4 060 8 0 100 1 2 0 v out ( v ) t emp ( c ) v out vs. temperature 4 0 45 5 0 55 60 6 5 4.5 5.5 6 .5 7 .5 8.5 9.5 10 .5 11 .5 1 2.5 13 .5 curr e nt l i m it (ma) v cc ( v ) current limit threshold vs. v cc voltage 0 2 4 6 8 10 1 2 1 4 16 1 8 2 0 4.5 5.5 6 .5 7 .5 8.5 9.5 10 .5 11 .5 1 2.5 13 .5 i nput c urrent (ma) v cc v oltage ( v ) v cc current vs. v cc voltage 0 5 10 1 5 2 0 25 30 3 5 4 0 45 5 0 4.5 5.5 6 .5 7 .5 8.5 9.5 10 .5 11 .5 1 2.5 13 .5 enable t ime ( sec) v cc ( v ) v oltage enable time vs. v cc voltage december 2005 4 m9999-120105
m ic 5 1 9 0 micrel functional characteristics ti me ( 100 s/div) output ( 10 m v /div) l o a d curr e nt (5a/div) input ( 100 m v /div) 10a load transient ti me ( 100 s/div) input ( 100 m v /div) output ( 10 m v /div) l o a d curr e nt (5a/div) transient response disable transient ti me ( 100 s/div) output (5 00 m v /div) e n able ( 1v /div) enable transient ti me ( 10 s/div) output (5 00 m v /div) e n able ( 1v /div) december 2005 5 m9999-120105
m ic 5 1 9 0 micrel functional diagram figure 1. MIC5190 block diagram functional description vin t he vin pin is connected to the n - c hannel drain. vin is the input power being supplied to the output. t his pin is also used to power the internal current limit comparator and compare the is e ns e voltage for current limit. t he voltage range is from 0 .9 v min to 5.5 v max. isense t he is e ns e pin is the other input to the current limit com- parator. t he output current is limited when the is e ns e pin's voltage is 5 0 m v less than the vin pin. i n cases where there is a current limited source and there isnt a need for current limit, this pin can be tied directly to vin . i ts operating voltage range, like the vin pin, is 0 .9 v min to 5.5 v max. vcc1, vcc2 vcc1 supplies the error amplifier and internal reference, while vcc 2 supplies the output gate drive. for this reason, ensure these pins have good input capacitor bypassing for better performance. t he operating range is from 4.5 v to 13 .2 v and both vcc pins should be tied together. ensure that the voltage supplied is greater than a gate-source threshold above the output voltage for the n - c hannel m os fe t se- lected. output t he output drives the external n - c hannel m os fe t and is powered from v cc . t he output can sink and source over 1 5 0 ma of current to drive either an n - c hannel m os fe t or an external npn transistor. t he output drive also has short circuit current protection. enable t he m ic 5 1 9 0 comes with an active-high enable pin that allows the regulator to be disabled. forcing the enable pin low disables the regulator and sends it into a low off-mode- current state. forcing the enable pin high enables the output voltage. t he enable pin cannot be left floating; a floating enable pin may cause an indeterminate state on the output. fb t he feedback pin is used to sense the output voltage for regulation. t he feedback pin is compared to an internal 0 .5 v reference and the output adjusts the gate voltage accordingly to maintain regulation. s ince the feedback biasing current is typically 13 a, smaller feedback resistors should be used to minimize output voltage error. comp co m p is the external compensation pin. t his allows com- plete control over the loop to allow stability for any type of output capacitor, load currents and output voltage. a detailed explanation of how to compensate the m ic 5 1 9 0 is in the d esigning with the m ic 5 1 9 0 section. sgnd, pgnd sgnd is the internal signal ground which provides an iso- lated ground path from the high current output driver. t he signal ground provides the grounding for noise sensitive circuits such as the current limit comparator, error amplifier and the internal reference voltage. pgnd is the power ground and is the grounding path for the output driver. output contro l a nd le v el s h i f t curr e nt l i m it am p l i f i e r e rror am p l i f i e r e n able vin is vcc 2 out pgnd fb 0 .5 v 5 0 m v sgnd e n vcc1 co m p int e rn al vo l t a g e r e gu la tor december 2005 6 m9999-120105
m ic 5 1 9 0 micrel applications information designing with the MIC5190 anatomy of a transient response t he measure of a regulator is how accurately and effectively it can maintain a set output voltage, regardless of the load's power demands. o ne measure of regulator response is the load step. t he load step gauges how the regulator responds to a change in load current. figure 2 is a look at the transient response to a load step. figure 2. typical transient response at the start of a circuit's power demand, the output voltage is regulated to its set point, while the load current runs at a constant rate. for many different reasons, a load may ask for more current without warning. when this happens, the regu- lator needs some time to determine the output voltage drop. t his is determined by the speed of the control loop. s o, until enough time has elapsed, the control loop is oblivious to the voltage change. t he output capacitor must bear the burden of maintaining the output voltage. s ince this is a sudden change in voltage, the capacitor will try to maintain voltage by discharging current to the output. t he first voltage drop is due to the output capacitor's e s l (equiva- lent series inductance). t he e s l will resist a sudden change in current from the capacitor and drop the voltage quickly. t he amount of voltage drop during this time will be proportional to the output capacitor's e s l and the speed at which the load steps. s lower load current transients will reduce this effect. p lacing multiple small capacitors with low e s l in parallel can help reduce the total e s l and reduce voltage droop during high speed transients. for high speed transients, the greatest voltage deviation will generally be caused by output capacitor e s l and parasitic inductance. after the current has overcome the effects of the e s l, the output voltage will begin to drop proportionally to time and inversely proportional to output capacitance. o utput voltage variation will depend on two factors: loop bandwidth and output capacitance. t he output capacitance will determine how far the voltage will fall over a given time. with more capacitance, the drop in voltage will fall at a decreased rate. t his is the reason that more capacitance provides a better transient response for the same given bandwidth. t he time it takes for the regulator to respond is directly proportional to its bandwidth gain. higher bandwidth control loops respond quicker causing a reduced drop on the supply for the same amount of capacitance. final recovery back to the regulated voltage is the final phase of transient response and the most important factors are gain and time. higher gain at higher frequency will get the output voltage closer to its regulation point quicker. t he final settling point will be determined by the load regulation, which is proportional to dc ( 0 hz) gain and the associated loss terms. t here are other factors that contribute to large signal tran- sient response, such as source impedance, phase margin, and psrr . for example, if the input voltage drops due to source impedance during a load transient, this will contribute to the output voltage deviation by filtering through to the output reduced by the loops psrr at the frequency of the voltage transient. i t is straightforward: good input capaci- tance reduces the source impedance at high frequencies. having between 3 5 and 45 of phase margin will help speed up the recovery time. t his is caused by the initial overshoot in response to the loop sensing a low voltage. compensation t he m ic 5 1 9 0 has the ability to externally control gain and bandwidth. t his allows the m ic 5 1 9 0 design to be individually tailored for different applications. i n designing the m ic 5 1 9 0 , it is important to maintain ad- equate phase margin. t his is generally achieved by having the gain cross the 0 db point with a single pole 2 0 db/decade roll-off. t he compensation pin is configured as figure 3 demonstrates. error amplifier d river 3 .42m ? 2 0 pf i nternal external c omp figure 3. internal compensation ? v l di dt = ? v c idt = 1 ? v c idt = 1 ? v c idt = 1 ? v l di dt = ? v l di dt = t ime idt c v = 1 bw 1 load c urrent o utput v oltage a c - c oupled o utput voltage vs. time during recovery is directly proportional to gain vs. frequency. ? v = l di dt december 2005 7 m9999-120105
m ic 5 1 9 0 micrel t his places a pole at 2. 3 khz at 8 0 db and calculates as follows. f mpf f khz p p = = 1 2 3 42 2 0 2 3 2 . . ? -2 0 0 2 0 4 0 60 8 0 100 0 . 01 0 . 1 1 10 100 1000 10000 100000 frequency (khz) gain ( db) -45 0 45 9 0 13 5 1 8 0 225 phase (deg) figure 4. internal compensation frequency response t here is single pole roll off. for most applications, an output capacitor is required. t he output capacitor and load resis- tance create another pole. t his causes a two-pole system and can potentially cause design instability with inadequate phase margin. external compensation is required. by provid- ing a dominant pole and zeroCallowing the output capacitor and load to provide the final poleCa net single pole roll off is created, with the zero canceling the dominant pole. figure 5 demonstrates placing an external capacitor ( c co m p ) and resistor ( r co m p ) for the external pole-zero combination. where the dominant pole can be calculated as follows: error amplifier d river 3 .42m ? 2 0 pf i nternal external c omp r co m p c co m p figure 5. external compensation f m c p co m p = 1 2 3 42 .? and the zero can be calculated as follows: f rc z co m pco m p = 1 2 t his allows for high dc gain, and high bandwidth with the output capacitor and the load providing the final pole. figure 6. external compensation frequency response i t is recommended that the gain bandwidth should be de- signed to be less than 1 mhz. t his is because most capaci- tors lose capacitance at high frequency and becoming resis- tive or inductive. t his can be difficult to compensate for and can create high frequency ringing or worse, oscillations. by increasing the amount of output capacitance, transient response can be improved in multiple ways. first, the rate of voltage drop vs. time is decreased. also, by increasing the output capacitor, the pole formed by the load and the output capacitor decreases in frequency. t his allows for the increas- ing of the compensation resistor, creating a higher mid-band gain. figure 7. increasing output capacitance t his will have the effect of both decreasing the voltage drop as well as returning closer and faster to the regulated voltage during the recovery time. mosfet selection t he typical pass element for the m ic 5 1 9 0 is an n - c hannel m os fe t . t here are multiple considerations when choosing a m os fe t . t hese include: ? v in to v out differential ? o utput current ? c ase size/thermal characteristics ? g ate capacitance ( c iss < 10 nf) ? g ate to source threshold -2 0 0 2 0 4 0 60 8 0 100 0 . 01 0 . 1 1 10 100 1000 10000 100000 frequency (khz) gain (db ) -45 0 45 9 0 13 5 1 8 0 225 phase (deg) t he d ominant p ole external zero r load c out p ole c comp m fp = 42 . 3 2 1 c comp r comp fz = 2 1 -2 0 0 2 0 4 0 60 8 0 100 0 . 01 0 . 1 1 10 100 1000 10000 100000 frequency (khz) gain (db ) -45 0 45 9 0 13 5 1 8 0 225 phase (deg) i ncreasing c out reduces the load resistance and output capacitor pole allowing for an increase in mid-band gain. december 2005 8 m9999-120105
m ic 5 1 9 0 micrel t he v in (min) to v out ratio and current will determine the maximum r dson required. for example, for a 1 .8 v ( 5%) to 1 .5 v conversion at 5a of load current, dropout voltage can be calculated as follows (using v in (min)): r vv i r 1 71v 1 5 v 5a r m dson in out out dson dson = ? () = ? () = .. 42 ? r unning the n - c hannel in dropout will seriously affect tran- sient response and psrr (power supply ripple rejection). for this reason, we want to select a m os fe t that has lower than 42m ? for our example application. s ize is another important consideration. most importantly, the design must be able to handle the amount of power being dissipated. t he amount of power dissipated can be calculated as follows (using v in (max)): p d = ( v in C v out ) i out p d = ( 1 .89 v C 1 .5 v ) 5a p d = 1 .95w n ow that we know the amount of power we will be dissipating, we will need to know the maximum ambient air temperature. for our case were going to assume a maximum of 6 5 c ambient temperature. d ifferent m os fe t s have different maximum operating junction temperatures. most m os fe t s are rated to 1 5 0 c , while others are rated as high as 17 5 c . i n this case, were going to limit our maximum junction temperature to 1 25 c . t he m ic 5 1 9 0 has no internal thermal protection for the m os fe t so it is important that the design provides margin for the maximum junction temperature. o ur design will maintain better than 1 25 c junction temperature with 1 .95w of power dissipation at an ambient temperature of 6 5 c . o ur thermal resistance calculates as follows: s o our package must have a thermal resistance less than 31 c /w. t able 1 . shows a good approximation of power dissipation and package recommendation. package power dissipation tsop - 6 <85 0 mw tssop -8 <95 0 mw tssop -8 < 1 w p ower p ak? 1 2 1 2-8 < 1 . 1 w so -8 < 1 . 1 25w p ower p ak? so -8 d - p ack < 1 .4w to -22 0 / to -2 63 ( d 2 p ack) > 1 .4w table 1. power dissipation and package recommendation i n our example, our power dissipation is greater than 1 .4w, so well choose a to -2 63 ( d 2 p ack) n - c hannel m os fe t . ja is calculated as follows. ja = j c + cs + s a where j c is the junction-to-case resistance, cs is the case-to-sink resistance and the s a is the sink-to-ambient air resistance. i n the d 2 package weve selected, the j c is 2 c /w. t he cs , assuming we are using the pc b as the heat sink, can be approximated to 0 .2 c /w. t his allows us to calculate the minimum s a : s a = ja C cs C j c s a = 31 c /w C 0 .2 c /w C 2 c /w s a = 28.8 c /w r eferring to application hint 17 , d esigning pc b heat s inks , the minimum amount of copper area for a d 2 p ack at 28.8 c /w is 2 7 5 0 mm 2 (or 0 .42 6 in 2 ). t he solid line denotes convection heating only (2 oz. copper) and the dotted line shows thermal resistance with 25 0 lfm air- flow. t he copper area can be significantly reduced by increasing airflow or by adding external heat sinks. figure 8. pc board heat sink another important characteristic is the amount of gate capacitance. large gate capacitance can reduce tran- sient performance by reducing the ability of the m ic 5 1 9 0 to slew the gate. i t is recommended that the m os fe t used has an input capacitance < 10 nf ( c iss ). ja jj d ja ja t max t ambient p 1 25 c6 5 c 1 .95w c w = () ? () = ? = 31 / pc board heat sink thermal resistance vs. area december 2005 9 m9999-120105
m ic 5 1 9 0 micrel t he gate-source threshold specified in most m os fe t data sheets refers to the minimum voltage needed to fully enhance the m os fe t . although for the most part, the m os fe t will be operating in the linear region and the v gs (gate-source voltage) will be less than the fully enhanced v gs , it is recommended the v cc voltage has 2 v over the minimum v gs and output voltage. t his is due to the saturation voltage of the m ic 5 1 9 0 output driver. v cc1 ,2 2 v + v gs + v out for our example, with a 1 .5 v output voltage, our m os fe t is fully enhanced at 4.5 v gs , and so our v cc voltage should be greater or equal to 8 v . input capacitor g ood input bypassing is important for improved perfor- mance. low e sr and low e s l input capacitors reduce both the drain of the n - c hannel m os fe t , as well as the source impedance to the m ic 5 1 9 0 . when a load transient on the output occurs, the load step will also appear on the input. d eviations on the input voltage will be reduced by the m ic 5 1 9 0 s psrr , but nonetheless appear on the output. t here really is no minimum input capacitance, but it is recommended that the input capacitance be equal to or greater than the output capacitance for best performance. output capacitor t he m ic 5 1 9 0 is stable with any type or value of output capacitor (even without any output capacitor!). t his allows the output capacitor to select which parameters of the regu- lator are important. i n cases where transient response is the most important, low e sr and low e s l ceramic capacitors are recommended. also, the more capacitance on the output, the better the transient response. 4 5 3 1 2 vin e n / uv l o cs h vout fb l sd 5 60 pf 8. 06 k b st co m p 6 h sd 1 2 vs w 11 gnd d1 sd103 bw s 2.2 f 10v 0 . 1 f 10 8 vdd 7 u1 mic2198-bml 9 100 pf 11 .5k 100 k v out v out cs h j2 e n j 1 + v in v in cs h v out 10 k 10 k 10 ? 10 ? 1 f 25 v 330 f 16v 10 f 10 f 10 f 10 f 10 f 22 f 1v out @ 10 a d 2 1n 58 1 9hw ir f 7 82 1 ir f 7 82 1 l 1 1 .8 h cd e p13 4- 1r 8m c -h 1 2.4k 330 f t antalum out vcc1 vcc 2 vin fb is e ns e co m p gnd 10 nf 100 ? 100 ? 10 ? MIC5190 1 f v out figure 9. post regulator december 2005 10 m 9999-120105
m ic 5 1 9 0 micrel feedback resistors ir3716s r1 cout r 2 fb gnd MIC5190 v out figure 10. adjustable output t he feedback resistors adjust the output to the desired voltage and can be calculated as follows: vv r1 r 2 out r ef =+ ? ? ? ? ? ? 1 v r ef is equal to 0 .5 v for the m ic 5 1 9 0 . t he minimum output voltage ( r1 = 0 ) is 0 .5 v . for output voltages greater than 1v , use the m ic 5 1 9 1 . t he resistor tolerance adds error to the output voltage. t hese errors are accumulative for both r1 and r 2. for example, our resistors selected have a 1 % tolerance. t his will contribute to a 2% additional error on the output voltage. t he feedback resistors must also be small enough to allow enough current to the feedback node. large feedback resis- tors will contribute to output voltage error. vr1i v1 k 1 a v m v e rror fb e rror e rror = = = ? 2 1 2 for our example application, this will cause an increase in output voltage of 1 2m v . for the percentage increase, v v v v 1 2m v 1 .5 v v e rror e rror out e rror e rror % % %.% = = = 100 100 0 8 by reducing r1 to 100 ? , the error contribution by the feed- back resistors and feedback current is reduced to less than 0 . 1 %. t his is the reason r1 should not be greater than 100 ? . applying the MIC5190 linear regulator t he primary purpose of the m ic 5 1 9 0 is as a linear regulator, which enables an input supply voltage to drop down through the resistance of the pass element to a regulated output voltage. active filter another application for the m ic 5 1 9 0 is as an active filter on the output of a switching regulator. t his improves the power supply in several ways. first, using the m ic 5 1 9 0 as a filter on the output can signifi- cantly reduce high frequency noise. s witching power sup- plies tends to create noise at the switching frequency in the form of a triangular voltage ripple. high frequency noise is also created by the high-speed switching transitions. a lot of time, effort , and money are thrown into the design of switching regulators to minimize these effects as much as possible. figure 9 shows the m ic 5 1 9 0 as a post regulator. figure 11. ripple reduction figure 11 shows the amount of ripple reduction for a 5 00 khz switching regulator. t he fundamental switching frequency is reduced from greater than 100 m v to less than 10 m v . figure 12. 10a load transient t he transient response also contributes to the overall a c output voltage deviation. figure 1 2 shows a 1 a to 10 a load transient. t he top trace is the output of the switching regulator (same circuit as figure 10 ). t he output voltage undershoots by 100 m v . just by their topology, linear regulators have the ability to respond at much higher speeds than a switching regulator. linear regulators do not have the limitation or restrictions of switching regulators which must reduce their bandwidth to less than their switching frequency. ti me ( 100 s/div) output ( 10 m v /div) l o a d curr e nt (5a/div) input ( 100 m v /div) ti me ( 1 s/div) input ripp le ( 100 m v /div) output ( 10 m v /div) v out = 1v i l o a d = 10 a december 2005 11 m 9999-120105
m ic 5 1 9 0 micrel u sing the m ic 5 1 9 0 as a filter for a switching regulator reduces output noise due to ripple and high frequency switch- ing noise. i t also reduces undershoot (figure 1 2) and over- shoot (figure 13 ) due to load transients with decreased capacitance. figure 13. transient response d ue to the high dc gain (8 0 db) of the m ic 5 1 9 0 , it also adds increased output accuracy and extremely high load regula- tion. distributed power supply as technology advances and processes move to smaller and smaller geometries, voltage requirements go down and cur- rent requirements go up. t his creates unique challenges when trying to supply power to multiple devices on a board. when there is one load to power, the difficulties are not quite as complex; trying to distribute power to multiple loads from one supply is much more problematic. i f a large circuit board has multiple small-geometry a sic s, it will require the powering of multiple loads with its one power source. assuming that the a sic s are dispersed throughout the board and that the core voltage requires a regulated 1v , figure 1 4 shows the long traces from the power supply to the a sic loads. n ot only do we have to contend with the toler- ance of the supply (line regulation, load regulation, output accuracy, and temperature tolerances), but the trace lengths create additional issues with resistance and inductance. with lower voltages these parasitic values can easily bump the output voltage out of a usable tolerance. load load s witching p ower s upply circuit board load load long t races figure 14. board layout but by placing multiple small m ic 5 1 9 0 circuits close to each load, the parasitic trace elements caused by distance to the power supply are almost completely negated. by adjusting the switching supply voltage to 1 .2 v , for our example, the m ic 5 1 9 0 will provide accurate 1v output, efficiently and with very little noise. figure 15. improved distributed supplies load load s witching p ower s upply circuit board load load m ic 5 1 9 0 m ic 5 1 9 0 m ic 5 1 9 0 m ic 5 1 9 0 ti me ( 100 s/div) input ( 100 m v /div) output ( 10 m v /div) l o a d curr e nt (5a/div) december 2005 12 m9999-120105
m ic 5 1 9 0 micrel micrel, inc. 1849 fortune drive san jose, ca 95131 usa t el + 1 (4 0 8) 944- 0 8 00 fax + 1 (4 0 8) 4 7 4- 1000 web http://www.micrel.com t he information furnished by micrel in this data sheet is believed to be accurate and reliable. however, no responsibility is as sumed by micrel for its use. micrel reserves the right to change circuitry and specifications at any time without notification to the customer. micrel p roducts are not designed or authorized for use as components in life support appliances, devices or systems where malfunction o f a product can reasonably be expected to result in personal injury. life support devices or systems are devices or systems that (a) are intend ed for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant inj ury to the user. a p urchasers use or sale of micrel p roducts for use in life support appliances, devices or systems is at p urchasers own risk and p urchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. ? 2 00 4 micrel, i ncorporated. 0 . 1 5 ( 0 . 006 ) 0 . 0 5 ( 0 . 00 2) 0 .5 0 b sc ( 0 . 0 2 0 ) 6 max 0 m in 3 . 1 5 ( 0 . 1 22) 2.85 ( 0 . 11 4) 3 . 10 ( 0 . 1 22) 2.9 0 ( 0 . 11 4) 0 . 30 ( 0 . 01 2) 0 . 1 5 ( 0 . 006 ) 0 .2 6 ( 0 . 010 ) 0 . 10 ( 0 . 00 4) 1 . 10 ( 0 . 0 4 3 ) 0 .94 ( 0 . 037 ) di me nsions : mm ( inc h) 0 . 70 ( 0 . 0 28) 0 .4 0 ( 0 . 016 ) 4.9 0 b sc ( 0 . 1 9 3 ) 10-pin ms0p (mm) 10-lead mlf (ml) december 2005 13 m 9999-120105

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